Methods for reducing near-field radiation and specific absorption rate (sar) values in communications devices

ABSTRACT

A method is provided introducing a phase offset between signals applied to antenna ports of an antenna having multiple antenna elements, such that a first signal applied to one of the antenna ports operatively coupled to one of the antenna elements has a different phase than a second signal applied to another one of antenna ports operatively coupled to another one of the antenna elements to obtain an antenna pattern control. A reduced power is used that is lower than the power used in a non-pattern control operation of the antenna such that a wireless link performance criteria is met with equipment at a far-field point using the reduced power compared to the non-pattern control operation, thereby reducing near field radiation.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of and claims priority to U.S. patentapplication Ser. No. 14/754,900 filed Jun. 30, 2015 entitled Methods ForReducing Near-Field Radiation And Specific Absorption Rate (SAR) Valuesin Communications Devices, which is a continuation of U.S. patentapplication Ser. No. 14/225,640 filed Mar. 26, 2014 entitled Methods ForReducing Near-Field Radiation And Specific Absorption Rate (SAR) Valuesin Communications Devices (issued as U.S. Pat. No. 9,100,096), which isa continuation of U.S. patent application Ser. No. 13/726,871 filed Dec.26, 2012 entitled Methods For Reducing Near-Field Radiation And SpecificAbsorption Rate (SAR) Values in Communications Devices (issued as U.S.Pat. No. 8,723,743), which is a continuation of U.S. patent applicationSer. No. 12/786,032 filed May 24, 2010 entitled Methods For ReducingNear-Field Radiation And Specific Absorption Rate (SAR) Values InCommunications Devices (issued as U.S. Pat. No. 8,344,956), which is acontinuation-in-part of U.S. patent application Ser. No. 12/750,196filed Mar. 30, 2010 entitled Multimode Antenna Structure (issued as U.S.Pat. No. 8,164,538), which is a continuation of U.S. patent applicationSer. No. 12/099,320 filed Apr. 8, 2008, entitled Multimode AntennaStructure (issued as U.S. Pat. No. 7,688,273), which is acontinuation-in-part of U.S. patent application Ser. No. 11/769,565filed Jun. 27, 2007 entitled Multimode Antenna Structure (issued as U.S.Pat. No. 7,688,275), which is a non-provisional of U.S. ProvisionalPatent Application No. 60/925,394 filed on Apr. 20, 2007 entitledMultimode Antenna Structure and U.S. Provisional Patent Application No.60/916,655 filed on May 8, 2007 also entitled Multimode AntennaStructure. U.S. patent application Ser. No. 12/786,032 is also anon-provisional of U.S. Provisional Patent Application No. 61/181,176filed on May 26, 2009 also entitled Multimode Antenna Structure. Each ofthe above-identified applications is incorporated by reference herein.

BACKGROUND

The present invention relates generally to wireless communicationsdevices and, more particularly, to methods for reducing near-fieldradiation and specific absorption rate (SAR) values in such devices.

Many communications devices have multiple antennas that are packagedclose together (e.g., less than a quarter of a wavelength apart) andthat can operate simultaneously within the same frequency band. Commonexamples of such communications devices include portable communicationsproducts such as cellular handsets, personal digital assistants (PDAs),and wireless networking devices or data cards for personal computers(PCs). Many system architectures (such as Multiple Input Multiple Output(MIMO)) and standard protocols for mobile wireless communicationsdevices (such as 802.11n for wireless LAN, and 3G data communicationssuch as 802.16e (WiMAX), HSDPA, and 1xEVDO) require multiple antennasoperating simultaneously.

BRIEF SUMMARY OF EMBODIMENTS OF THE INVENTION

In accordance with one or more embodiments, a method is provided forreducing near-field radiation and specific absorption rate (SAR) valuesin a communications device. The communications device includes amultimode antenna structure transmitting and receiving electromagneticsignals and circuitry for processing signals communicated to and fromthe antenna structure. The antenna structure comprises: a plurality ofantenna ports operatively coupled to the circuitry; a plurality ofantenna elements, each operatively coupled to a different one of theantenna ports; and one or more connecting elements electricallyconnecting the antenna elements at a location on each antenna elementthat is spaced apart from an antenna port coupled thereto to form asingle radiating structure and such that electrical currents on oneantenna element flow to a connected neighboring antenna element andgenerally bypass the antenna port coupled to the neighboring antennaelement, the electrical currents flowing through the one antenna elementand the neighboring antenna element being generally equal in magnitude,such that an antenna mode excited by one antenna port is generallyelectrically isolated from a mode excited by another antenna port at agiven desired signal frequency range and the antenna structure generatesdiverse antenna patterns. The method includes adjusting the relativephase between signals fed to neighboring antenna ports of the antennastructure such that a signal fed to the one antenna port has a differentphase than a signal fed to the neighboring antenna port to provideantenna pattern control and to increase gain in a selected directiontoward a receive point. The method features using a transmit power lowerthan the transmit power used in a non-pattern control operation of theantenna structure such that the communications device obtains generallyequivalent wireless link performance with the receive point usingreduced transmit power compared to the non-pattern control operation,thereby reducing the specific absorption rate.

In accordance with one or more further embodiments, a method is providedfor reducing near-field radiation and specific absorption rate (SAR)values in a communications device. The communications device includes anantenna array for transmitting and receiving electromagnetic signals andcircuitry for processing signals communicated to and from the antennaarray. The antenna array comprises a plurality of radiating elementseach having an antenna port operatively coupled to the circuitry. Themethod includes adjusting the relative phase between signals fed to theantenna ports of the antenna array such that a signal fed to one antennaport has a different phase than a signal fed to another antenna port toprovide antenna pattern control and to increase gain in a selecteddirection toward a receive point. The method features using a transmitpower lower than the transmit power used in a non-pattern controloperation of the antenna array such that the communications deviceobtains generally equivalent wireless link performance with the receivepoint using reduced transmit power compared to the non-pattern controloperation, thereby reducing the specific absorption rate.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A illustrates an antenna structure with two parallel dipoles.

FIG. 1B illustrates current flow resulting from excitation of one dipolein the antenna structure of FIG. 1A.

FIG. 1C illustrates a model corresponding to the antenna structure ofFIG. 1A.

FIG. 1D is a graph illustrating scattering parameters for the FIG. 1Cantenna structure.

FIG. 1E is a graph illustrating the current ratios for the FIG. 1Cantenna structure.

FIG. 1F is a graph illustrating gain patterns for the FIG. 1C antennastructure.

FIG. 1G is a graph illustrating envelope correlation for the FIG. 1Cantenna structure.

FIG. 2A illustrates an antenna structure with two parallel dipolesconnected by connecting elements in accordance with one or moreembodiments of the invention.

FIG. 2B illustrates a model corresponding to the antenna structure ofFIG. 2A.

FIG. 2C is a graph illustrating scattering parameters for the FIG. 2Bantenna structure.

FIG. 2D is a graph illustrating scattering parameters for the FIG. 2Bantenna structure with lumped element impedance matching at both ports.

FIG. 2E is a graph illustrating the current ratios for the FIG. 2Bantenna structure.

FIG. 2F is a graph illustrating gain patterns for the FIG. 2B antennastructure.

FIG. 2G is a graph illustrating envelope correlation for the FIG. 2Bantenna structure.

FIG. 3A illustrates an antenna structure with two parallel dipolesconnected by meandered connecting elements in accordance with one ormore embodiments of the invention.

FIG. 3B is a graph showing scattering parameters for the FIG. 3A antennastructure.

FIG. 3C is a graph illustrating current ratios for the FIG. 3A antennastructure.

FIG. 3D is a graph illustrating gain patterns for the FIG. 3A antennastructure.

FIG. 3E is a graph illustrating envelope correlation for the FIG. 3Aantenna structure.

FIG. 4 illustrates an antenna structure with a ground or counterpoise inaccordance with one or more embodiments of the invention.

FIG. 5 illustrates a balanced antenna structure in accordance with oneor more embodiments of the invention.

FIG. 6A illustrates an antenna structure in accordance with one or moreembodiments of the invention.

FIG. 6B is a graph showing scattering parameters for the FIG. 6A antennastructure for a particular dipole width dimension.

FIG. 6C is a graph showing scattering parameters for the FIG. 6A antennastructure for another dipole width dimension.

FIG. 7 illustrates an antenna structure fabricated on a printed circuitboard in accordance with one or more embodiments of the invention.

FIG. 8A illustrates an antenna structure having dual resonance inaccordance with one or more embodiments of the invention.

FIG. 8B is a graph illustrating scattering parameters for the FIG. 8Aantenna structure.

FIG. 9 illustrates a tunable antenna structure in accordance with one ormore embodiments of the invention.

FIGS. 10A and 10B illustrate antenna structures having connectingelements positioned at different locations along the length of theantenna elements in accordance with one or more embodiments of theinvention.

FIGS. 10C and 10D are graphs illustrating scattering parameters for theFIGS. 10A and 10B antenna structures, respectively.

FIG. 11 illustrates an antenna structure including connecting elementshaving switches in accordance with one or more embodiments of theinvention.

FIG. 12 illustrates an antenna structure having a connecting elementwith a filter coupled thereto in accordance with one or more embodimentsof the invention.

FIG. 13 illustrates an antenna structure having two connecting elementswith filters coupled thereto in accordance with one or more embodimentsof the invention.

FIG. 14 illustrates an antenna structure having a tunable connectingelement in accordance with one or more embodiments of the invention.

FIG. 15 illustrates an antenna structure mounted on a PCB assembly inaccordance with one or more embodiments of the invention.

FIG. 16 illustrates another antenna structure mounted on a PCB assemblyin accordance with one or more embodiments of the invention.

FIG. 17 illustrates an alternate antenna structure that can be mountedon a PCB assembly in accordance with one or more embodiments of theinvention.

FIG. 18A illustrates a three mode antenna structure in accordance withone or more embodiments of the invention.

FIG. 18B is a graph illustrating the gain patterns for the FIG. 18Aantenna structure.

FIG. 19 illustrates an antenna and power amplifier combiner applicationfor an antenna structure in accordance with one or more embodiments ofthe invention.

FIGS. 20A and 20B illustrate a multimode antenna structure useable,e.g., in a WiMAX USB or ExpressCard/34 device in accordance with one ormore further embodiments of the invention.

FIG. 20C illustrates a test assembly used to measure the performance ofthe antenna of FIGS. 20A and 20B.

FIGS. 20D to 20J illustrate test measurement results for the antenna ofFIGS. 20A and 20B.

FIGS. 21A and 21B illustrate a multimode antenna structure useable,e.g., in a WiMAX USB dongle in accordance with one or more alternateembodiments of the invention.

FIGS. 22A and 22B illustrate a multimode antenna structure useable,e.g., in a WiMAX USB dongle in accordance with one or more alternateembodiments of the invention.

FIG. 23A illustrates a test assembly used to measure the performance ofthe antenna of FIGS. 21A and 21B.

FIGS. 23B to 23K illustrate test measurement results for the antenna ofFIGS. 21A and 21B.

FIG. 24 is a schematic block diagram of an antenna structure with a beamsteering mechanism in accordance with one or more embodiments of theinvention.

FIGS. 25A to 25G illustrate test measurement results for the antenna ofFIG. 25A.

FIG. 26 illustrates the gain advantage of an antenna structure inaccordance with one or more embodiments of the invention as a functionof the phase angle difference between feedpoints.

FIG. 27A is a schematic diagram illustrating a simple dual-band branchline monopole antenna structure.

FIG. 27B illustrates current distribution in the FIG. 27A antennastructure.

FIG. 27C is a schematic diagram illustrating a spurline band stopfilter.

FIGS. 27D and 27E are test results illustrating frequency rejection inthe FIG. 27A antenna structure.

FIG. 28 is a schematic diagram illustrating an antenna structure with aband-rejection slot in accordance with one or more embodiments of theinvention.

FIG. 29A illustrates an alternate antenna structure with aband-rejection slot in accordance with one or more embodiments of theinvention.

FIGS. 29B and 29C illustrate test measurement results for the FIG. 29Aantenna structure.

FIG. 30 illustrates an exemplary USB dongle with two port antennastructure for pattern control application in the 1900 MHz band.

FIG. 31 illustrates SAR values as determined by simulation for thedevice of FIG. 30.

DETAILED DESCRIPTION

In accordance with various embodiments of the invention, multimodeantenna structures are provided for transmitting and receivingelectromagnetic signals in communications devices. The communicationsdevices include circuitry for processing signals communicated to andfrom an antenna structure. The antenna structure includes a plurality ofantenna ports operatively coupled to the circuitry and a plurality ofantenna elements, each operatively coupled to a different antenna port.The antenna structure also includes one or more connecting elementselectrically connecting the antenna elements such that an antenna modeexcited by one antenna port is generally electrically isolated from amode excited by another antenna port at a given signal frequency range.In addition, the antenna patterns created by the ports exhibitwell-defined pattern diversity with low correlation.

Antenna structures in accordance with various embodiments of theinvention are particularly useful in communications devices that requiremultiple antennas to be packaged close together (e.g., less than aquarter of a wavelength apart), including in devices where more than oneantenna is used simultaneously and particularly within the samefrequency band. Common examples of such devices in which the antennastructures can be used include portable communications products such ascellular handsets, PDAs, and wireless networking devices or data cardsfor PCs. The antenna structures are also particularly useful with systemarchitectures such as MIMO and standard protocols for mobile wirelesscommunications devices (such as 802.11n for wireless LAN, and 3G datacommunications such as 802.16e (WiMAX), HSDPA and 1xEVDO) that requiremultiple antennas operating simultaneously.

FIGS. 1A-1G illustrate the operation of an antenna structure 100. FIG.1A schematically illustrates the antenna structure 100 having twoparallel antennas, in particular parallel dipoles 102, 104, of length L.The dipoles 102, 104 are separated by a distance d, and are notconnected by any connecting element. The dipoles 102, 104 have afundamental resonant frequency that corresponds approximately to L=λ/2.Each dipole is connected to an independent transmit/receive system,which can operate at the same frequency. This system connection can havethe same characteristic impedance Z₀ for both antennas, which in thisexample is 50 ohms.

When one dipole is transmitting a signal, some of the signal beingtransmitted by the dipole will be coupled directly into the neighboringdipole. The maximum amount of coupling generally occurs near thehalf-wave resonant frequency of the individual dipole and increases asthe separation distance d is made smaller. For example, for d<λ/3, themagnitude of coupling is greater than 0.1 or −10 dB, and for d<λ/8, themagnitude of the coupling is greater than −5 dB.

It is desirable to have no coupling (i.e., complete isolation) or toreduce the coupling between the antennas. If the coupling is, e.g., −10dB, 10 percent of the transmit power is lost due to that amount of powerbeing directly coupled into the neighboring antenna. There may also bedetrimental system effects such as saturation or desensitization of areceiver connected to the neighboring antenna or degradation of theperformance of a transmitter connected to the neighboring antenna.Currents induced on the neighboring antenna distort the gain patterncompared to that generated by an individual dipole. This effect is knownto reduce the correlation between the gain patterns produced by thedipoles. Thus, while coupling may provide some pattern diversity, it hasdetrimental system impacts as described above.

Because of the close coupling, the antennas do not act independently andcan be considered an antenna system having two pairs of terminals orports that correspond to two different gain patterns. Use of either portinvolves substantially the entire structure including both dipoles. Theparasitic excitation of the neighboring dipole enables diversity to beachieved at close dipole spacing, but currents excited on the dipolepass through the source impedance, and therefore manifest mutualcoupling between ports.

FIG. 1C illustrates a model dipole pair corresponding to the antennastructure 100 shown in FIG. 1 used for simulations. In this example, thedipoles 102, 104 have a square cross section of 1 mm ×1 mm and length(L) of 56 mm. These dimensions yield a center resonant frequency of 2.45GHz when attached to a 50-ohm source. The free-space wavelength at thisfrequency is 122 mm. A plot of the scattering parameters S11 and S12 fora separation distance (d) of 10 mm, or approximately λ/12, is shown inFIG. 1D. Due to symmetry and reciprocity, S22=S11 and S12=S21. Forsimplicity, only S11 and S12 are shown and discussed. In thisconfiguration, the coupling between dipoles as represented by S12reaches a maximum of −3.7 dB.

FIG. 1E shows the ratio (identified as “Magnitude I2/I1” in the figure)of the vertical current on dipole 104 of the antenna structure to thaton dipole 102 under the condition in which port 106 is excited and port108 is passively terminated. The frequency at which the ratio ofcurrents (dipole 104/dipole 102) is a maximum corresponds to thefrequency of 180 degree phase differential between the dipole currentsand is just slightly higher in frequency than the point of maximumcoupling shown in FIG. 1D.

FIG. 1F shows azimuthal gain patterns for several frequencies withexcitation of port 106. The patterns are not uniformly omni-directionaland change with frequency due to the changing magnitude and phase of thecoupling. Due to symmetry, the patterns resulting from excitation ofport 108 would be the mirror image of those for port 106. Therefore, themore asymmetrical the pattern is from left to right, the more diversethe patterns are in terms of gain magnitude.

Calculation of the correlation coefficient between patterns provides aquantitative characterization of the pattern diversity. FIG. 1G showsthe calculated correlation between port 106 and port 108 antennapatterns. The correlation is much lower than is predicted by Clark'smodel for ideal dipoles. This is due to the differences in the patternsintroduced by the mutual coupling.

FIGS. 2A-2F illustrate the operation of an exemplary two port antennastructure 200 in accordance with one or more embodiments of theinvention. The two port antenna structure 200 includes twoclosely-spaced resonant antenna elements 202, 204 and provides both lowpattern correlation and low coupling between ports 206, 208. FIG. 2Aschematically illustrates the two port antenna structure 200. Thisstructure is similar to the antenna structure 100 comprising the pair ofdipoles shown in FIG. 1B, but additionally includes horizontalconductive connecting elements 210, 212 between the dipoles on eitherside of the ports 206, 208. The two ports 206, 208 are located in thesame locations as with the FIG. 1 antenna structure. When one port isexcited, the combined structure exhibits a resonance similar to that ofthe unattached pair of dipoles, but with a significant reduction incoupling and an increase in pattern diversity.

An exemplary model of the antenna structure 200 with a 10 mm dipoleseparation is shown in FIG. 2B. This structure has generally the samegeometry as the antenna structure 100 shown in FIG. 1C, but with theaddition of the two horizontal connecting elements 210, 212 electricallyconnecting the antenna elements slightly above and below the ports. Thisstructure shows a strong resonance at the same frequency as unattacheddipoles, but with very different scattering parameters as shown in FIG.2C. There is a deep drop-out in coupling, below −20 dB, and a shift inthe input impedance as indicated by S11. In this example, the bestimpedance match (S11 minimum) does not coincide with the lowest coupling(S12 minimum). A matching network can be used to improve the inputimpedance match and still achieve very low coupling as shown in FIG. 2D.In this example, a lumped element matching network comprising a seriesinductor followed by a shunt capacitor was added between each port andthe structure.

FIG. 2E shows the ratio (indicated as “Magnitude I2/I1” in the figure)of the current on dipole element 204 to that on dipole element 202resulting from excitation of port 206. This plot shows that below theresonant frequency, the currents are actually greater on dipole element204. Near resonance, the currents on dipole element 204 begin todecrease relative to those on dipole element 202 with increasingfrequency. The point of minimum coupling (2.44 GHz in this case) occursnear the frequency where currents on both dipole elements are generallyequal in magnitude. At this frequency, the phase of the currents ondipole element 204 lag those of dipole element 202 by approximately 160degrees.

Unlike the FIG. 1C dipoles without connecting elements, the currents onantenna element 204 of the FIG. 2B combined antenna structure 200 arenot forced to pass through the terminal impedance of port 208. Instead aresonant mode is produced where the current flows down antenna element204, across the connecting element 210, 212, and up antenna element 202as indicated by the arrows shown on FIG. 2A. (Note that this currentflow is representative of one half of the resonant cycle; during theother half, the current directions are reversed). The resonant mode ofthe combined structure features the following: (1) the currents onantenna element 204 largely bypass port 208, thereby allowing for highisolation between the ports 206, 208, and (2) the magnitude of thecurrents on both antenna elements 202, 204 are approximately equal,which allows for dissimilar and uncorrelated gain patterns as describedin further detail below.

Because the magnitude of currents is nearly equal on the antennaelements, a much more directional pattern is produced (as shown on FIG.2F) than in the case of the FIG. 1C antenna structure 100 withunattached dipoles. When the currents are equal, the condition fornulling the pattern in the x (or phi=0) direction is for the phase ofcurrents on dipole 204 to lag those of dipole 202 by the quantity π−kd(where k=2π/λ, and λ is the effective wavelength). Under this condition,fields propagating in the phi=0 direction from dipole 204 will be 180degrees out of phase with those of dipole 202, and the combination ofthe two will therefore have a null in the phi=0 direction.

In the model example of FIG. 2B, d is 10 mm or an effective electricallength of λ/12. In this case, kd equates π/6 or 30 degrees, and so thecondition for a directional azimuthal radiation pattern with a nulltowards phi=0 and maximum gain towards phi=180 is for the current ondipole 204 to lag those on dipole 202 by 150 degrees. At resonance, thecurrents pass close to this condition (as shown in FIG. 2E), whichexplains the directionality of the patterns. In the case of theexcitation of dipole 204, the radiation patterns are the mirror oppositeof those of FIG. 2F, and maximum gain is in the phi=0 direction. Thedifference in antenna patterns produced from the two ports has anassociated low predicted envelope correlation as shown on FIG. 2G. Thusthe combined antenna structure has two ports that are isolated from eachother and produce gain patterns of low correlation.

Accordingly, the frequency response of the coupling is dependent on thecharacteristics of the connecting elements 210, 212, including theirimpedance and electrical length. In accordance with one or moreembodiments of the invention, the frequency or bandwidth over which adesired amount of isolation can be maintained is controlled byappropriately configuring the connecting elements. One way to configurethe cross connection is to change the physical length of the connectingelement. An example of this is shown by the multimode antenna structure300 of FIG. 3A where a meander has been added to the cross connectionpath of the connecting elements 310, 312. This has the general effect ofincreasing both the electrical length and the impedance of theconnection between the two antenna elements 302, 304. Performancecharacteristics of this structure including scattering parameters,current ratios, gain patterns, and pattern correlation are shown onFIGS. 3B, 3C, 3D, and 3E, respectively. In this embodiment, the changein physical length has not significantly altered the resonant frequencyof the structure, but there is a significant change in S12, with largerbandwidth and a greater minimum value than in structures without themeander. Thus, it is possible to optimize or improve the isolationperformance by altering the electrical characteristic of the connectingelements.

Exemplary multimode antenna structures in accordance with variousembodiments of the invention can be designed to be excited from a groundor counterpoise 402 (as shown by antenna structure 400 in FIG. 4), or asa balanced structure (as shown by antenna structure 500 in FIG. 5). Ineither case, each antenna structure includes two or more antennaelements (402, 404 in FIGS. 4, and 502, 504 in FIG. 5) and one or moreelectrically conductive connecting elements (406 in FIGS. 4, and 506,508 in FIG. 5). For ease of illustration, only a two-port structure isillustrated in the example diagrams. However, it is possible to extendthe structure to include more than two ports in accordance with variousembodiments of the invention. A signal connection to the antennastructure, or port (418, 412 in FIGS. 4 and 510, 512 in FIG. 5), isprovided at each antenna element. The connecting element provideselectrical connection between the two antenna elements at the frequencyor frequency range of interest. Although the antenna is physically andelectrically one structure, its operation can be explained byconsidering it as two independent antennas. For antenna structures notincluding a connecting element such as antenna structure 100, port 106of that structure can be said to be connected to antenna 102, and port108 can be said to be connected to antenna 104. However, in the case ofthis combined structure such as antenna structure 400, port 418 can bereferred to as being associated with one antenna mode, and port 412 canbe referred to as being associated with another antenna mode.

The antenna elements are designed to be resonant at the desiredfrequency or frequency range of operation. The lowest order resonanceoccurs when an antenna element has an electrical length of one quarterof a wavelength. Thus, a simple element design is a quarter-wavemonopole in the case of an unbalanced configuration. It is also possibleto use higher order modes. For example, a structure formed fromquarter-wave monopoles also exhibits dual mode antenna performance withhigh isolation at a frequency of three times the fundamental frequency.Thus, higher order modes may be exploited to create a multiband antenna.Similarly, in a balanced configuration, the antenna elements can becomplementary quarter-wave elements as in a half-wave center-fed dipole.However, the antenna structure can also be formed from other types ofantenna elements that are resonant at the desired frequency or frequencyrange. Other possible antenna element configurations include, but arenot limited to, helical coils, wideband planar shapes, chip antennas,meandered shapes, loops, and inductively shunted forms such as PlanarInverted-F Antennas (PIFAs).

The antenna elements of an antenna structure in accordance with one ormore embodiments of the invention need not have the same geometry or bethe same type of antenna element. The antenna elements should each haveresonance at the desired frequency or frequency range of operation.

In accordance with one or more embodiments of the invention, the antennaelements of an antenna structure have the same geometry. This isgenerally desirable for design simplicity, especially when the antennaperformance requirements are the same for connection to either port.

The bandwidth and resonant frequencies of the combined antenna structurecan be controlled by the bandwidth and resonance frequencies of theantenna elements. Thus, broader bandwidth elements can be used toproduce a broader bandwidth for the modes of the combined structure asillustrated, e.g., in FIGS. 6A, 6B, and 6C. FIG. 6A illustrates amultimode antenna structure 600 including two dipoles 602, 604 connectedby connecting elements 606, 608. The dipoles 602, 604 each have a width(W) and a length (L) and are spaced apart by a distance (d). FIG. 6Billustrates the scattering parameters for the structure having exemplarydimensions: W=1 mm, L=57.2 mm, and d=10 mm. FIG. 6C illustrates thescattering parameters for the structure having exemplary dimensions:W=10 mm, L=50.4 mm, and d=10 mm. As shown, increasing W from 1 mm to 10mm, while keeping the other dimensions generally the same, results in abroader isolation bandwidth and impedance bandwidth for the antennastructure.

It has also been found that increasing the separation between theantenna elements increases the isolation bandwidth and the impedancebandwidth for an antenna structure.

In general, the connecting element is in the high-current region of thecombined resonant structure. It is therefore preferable for theconnecting element to have a high conductivity.

The ports are located at the feed points of the antenna elements as theywould be if they were operated as separate antennas. Matching elementsor structures may be used to match the port impedance to the desiredsystem impedance.

In accordance with one or more embodiments of the invention, themultimode antenna structure can be a planar structure incorporated,e.g., into a printed circuit board, as shown as FIG. 7. In this example,the antenna structure 700 includes antenna elements 702, 704 connectedby a connecting element 706 at ports 708, 710. The antenna structure isfabricated on a printed circuit board substrate 712. The antennaelements shown in the figure are simple quarter-wave monopoles. However,the antenna elements can be any geometry that yields an equivalenteffective electrical length.

In accordance with one or more embodiments of the invention, antennaelements with dual resonant frequencies can be used to produce acombined antenna structure with dual resonant frequencies and hence dualoperating frequencies. FIG. 8A shows an exemplary model of a multimodedipole structure 800 where the dipole antenna elements 802, 804 aresplit into two fingers 806, 808 and 810, 812, respectively, of unequallength. The dipole antenna elements have resonant frequencies associatedwith each the two different finger lengths and accordingly exhibit adual resonance. Similarly, the multimode antenna structure usingdual-resonant dipole arms exhibits two frequency bands where highisolation (or small S21) is obtained as shown in FIG. 8B.

In accordance with one or more embodiments of the invention, a multimodeantenna structure 900 shown in FIG. 9 is provided having variable lengthantenna elements 902, 904 forming a tunable antenna. This may be done bychanging the effective electrical length of the antenna elements by acontrollable device such as an RF switch 906, 908 at each antennaelement 902, 904. In this example, the switch may be opened (byoperating the controllable device) to create a shorter electrical length(for higher frequency operation) or closed to create a longer electricallength (for lower frequency of operation). The operating frequency bandfor the antenna structure 900, including the feature of high isolation,can be tuned by tuning both antenna elements in concert. This approachmay be used with a variety of methods of changing the effectiveelectrical length of the antenna elements including, e.g., using acontrollable dielectric material, loading the antenna elements with avariable capacitor such as a MEMs device, varactor, or tunabledielectric capacitor, and switching on or off parasitic elements.

In accordance with one or more embodiments of the invention, theconnecting element or elements provide an electrical connection betweenthe antenna elements with an electrical length approximately equal tothe electrical distance between the elements. Under this condition, andwhen the connecting elements are attached at the port ends of theantenna elements, the ports are isolated at a frequency near theresonance frequency of the antenna elements. This arrangement canproduce nearly perfect isolation at particular frequency.

Alternately, as previously discussed, the electrical length of theconnecting element may be increased to expand the bandwidth over whichisolation exceeds a particular value. For example, a straight connectionbetween antenna elements may produce a minimum S21 of −25 dB at aparticular frequency and the bandwidth for which S21<−10 dB may be 100MHz. By increasing the electrical length, a new response can be obtainedwhere the minimum S21 is increased to −15 dB but the bandwidth for whichS21<−10 dB may be increased to 150 MHz.

Various other multimode antenna structures in accordance with one ormore embodiments of the invention are possible. For example, theconnecting element can have a varied geometry or can be constructed toinclude components to vary the properties of the antenna structure.These components can include, e.g., passive inductor and capacitorelements, resonator or filter structures, or active components such asphase shifters.

In accordance with one or more embodiments of the invention, theposition of the connecting element along the length of the antennaelements can be varied to adjust the properties of the antennastructure. The frequency band over which the ports are isolated can beshifted upward in frequency by moving the point of attachment of theconnecting element on the antenna elements away from the ports andtowards the distal end of the antenna elements. FIGS. 10A and 10Billustrate multimode antenna structures 1000, 1002, respectively, eachhaving a connecting element electrically connected to the antennaelements. In the FIG. 10A antenna structure 1000, the connecting element1004 is located in the structure such the gap between the connectingelement 1004 and the top edge of the ground plane 1006 is 3 mm. FIG. 10Cshows the scattering parameters for the structure showing that highisolation is obtained at a frequency of 1.15 GHz in this configuration.A shunt capacitor/series inductor matching network is used to providethe impedance match at 1.15 GHz. FIG. 10D shows the scatteringparameters for the structure 1002 of FIG. 10B, where the gap between theconnecting element 1008 and the top edge 1010 of the ground plane is 19mm. The antenna structure 1002 of FIG. 10B exhibits an operating bandwith high isolation at approximately 1.50 GHz.

FIG. 11 schematically illustrates a multimode antenna structure 1100 inaccordance with one or more further embodiments of the invention. Theantenna structure 1100 includes two or more connecting elements 1102,1104, each of which electrically connects the antenna elements 1106,1108. (For ease of illustration, only two connecting elements are shownin the figure. It should be understood that use of more than twoconnecting elements is also contemplated.) The connecting elements 1102,1104 are spaced apart from each other along the antenna elements 1106,1108. Each of the connecting elements 1102, 1104 includes a switch 1112,1110. Peak isolation frequencies can be selected by controlling theswitches 1110, 1112. For example, a frequency f1 can be selected byclosing switch 1110 and opening switch 1112. A different frequency f2can be selected by closing switch 1112 and opening switch 1110.

FIG. 12 illustrates a multimode antenna structure 1200 in accordancewith one or more alternate embodiments of the invention. The antennastructure 1200 includes a connecting element 1202 having a filter 1204operatively coupled thereto. The filter 1204 can be a low pass or bandpass filter selected such that the connecting element connection betweenthe antenna elements 1206, 1208 is only effective within the desiredfrequency band, such as the high isolation resonance frequency. Athigher frequencies, the structure will function as two separate antennaelements that are not coupled by the electrically conductive connectingelement, which is open circuited.

FIG. 13 illustrates a multimode antenna structure 1300 in accordancewith one or more alternate embodiments of the invention. The antennastructure 1300 includes two or more connecting elements 1302, 1304,which include filters 1306, 1308, respectively. (For ease ofillustration, only two connecting elements are shown in the figure. Itshould be understood that use of more than two connecting elements isalso contemplated.) In one possible embodiment, the antenna structure1300 has a low pass filter 1308 on the connecting element 1304 (which iscloser to the antenna ports) and a high pass filter 1306 on theconnecting element 1302 in order to create an antenna structure with twofrequency bands of high isolation, i.e., a dual band structure.

FIG. 14 illustrates a multimode antenna structure 1400 in accordancewith one or more alternate embodiments of the invention. The antennastructure 1400 includes one or more connecting elements 1402 having atunable element 1406 operatively connected thereto. The antennastructure 1400 also includes antenna elements 1408, 1410. The tunableelement 1406 alters the delay or phase of the electrical connection orchanges the reactive impedance of the electrical connection. Themagnitude of the scattering parameters S21/S12 and a frequency responseare affected by the change in electrical delay or impedance and so anantenna structure can be adapted or generally optimized for isolation atspecific frequencies using the tunable element 1406.

FIG. 15 illustrates a multimode antenna structure 1500 in accordancewith one or more alternate embodiments of the invention. The multimodeantenna structure 1500 can be used, e.g., in a WIMAX USB dongle. Theantenna structure 1500 can be configured for operation, e.g., in WiMAXbands from 2300 to 2700 MHz.

The antenna structure 1500 includes two antenna elements 1502, 1504connected by a conductive connecting element 1506. The antenna elementsinclude slots to increase the electrical length of the elements toobtain the desired operating frequency range. In this example, theantenna structure is optimized for a center frequency of 2350 MHz. Thelength of the slots can be reduced to obtain higher center frequencies.The antenna structure is mounted on a printed circuit board assembly1508. A two-component lumped element match is provided at each antennafeed.

The antenna structure 1500 can be manufactured, e.g., by metal stamping.It can be made, e.g., from 0.2 mm thick copper alloy sheet. The antennastructure 1500 includes a pickup feature 1510 on the connecting elementat the center of mass of the structure, which can be used in anautomated pick-and-place assembly process. The antenna structure is alsocompatible with surface-mount reflow assembly.

FIG. 16 illustrates a multimode antenna structure 1600 in accordancewith one or more alternate embodiments of the invention. As with antennastructure 1500 of FIG. 15, the antenna structure 1600 can also be used,e.g., in a WIMAX USB dongle. The antenna structure can be configured foroperation, e.g., in WiMAX bands from 2300 to 2700 MHz.

The antenna structure 1600 includes two antenna elements 1602, 1604,each comprising a meandered monopole. The length of the meanderdetermines the center frequency. The exemplary design shown in thefigure is optimized for a center frequency of 2350 MHz. To obtain highercenter frequencies, the length of the meander can be reduced.

A connecting element 1606 electrically connects the antenna elements. Atwo-component lumped element match is provided at each antenna feed.

The antenna structure can be fabricated, e.g., from copper as a flexibleprinted circuit (FPC) mounted on a plastic carrier 1608. The antennastructure can be created by the metalized portions of the FPC. Theplastic carrier provides mechanical support and facilitates mounting toa PCB assembly 1610. Alternatively, the antenna structure can be formedfrom sheet-metal.

FIG. 17 illustrates a multimode antenna structure 1700 in accordancewith another embodiment of the invention. This antenna design can beused, e.g., for USB, Express 34, and Express 54 data card formats. Theexemplary antenna structure shown in the figure is designed to operateat frequencies from 2.3 to 6 GHz. The antenna structure can befabricated, e.g., from sheet-metal or by FPC over a plastic carrier1702.

FIG. 18A illustrates a multimode antenna structure 1800 in accordancewith another embodiment of the invention. The antenna structure 1800comprises a three mode antenna with three ports. In this structure,three monopole antenna elements 1802, 1804, 1806 are connected using aconnecting element 1808 comprising a conductive ring that connectsneighboring antenna elements. The antenna elements are balanced by acommon counterpoise, or sleeve 1810, which is a single hollow conductivecylinder. The antenna has three coaxial cables 1812, 1814, 1816 forconnection of the antenna structure to a communications device. Thecoaxial cables 1812, 1814, 1816 pass through the hollow interior of thesleeve 1810. The antenna assembly may be constructed from a singleflexible printed circuit wrapped into a cylinder and may be packaged ina cylindrical plastic enclosure to provide a single antenna assemblythat takes the place of three separate antennas. In one exemplaryarrangement, the diameter of the cylinder is 10 mm and the overalllength of the antenna is 56 mm so as to operate with high isolationbetween ports at 2.45 GHz. This antenna structure can be used, e.g.,with multiple antenna radio systems such as MIMO or 802.11N systemsoperating in the 2.4 to 2.5 GHz bands. In addition to port to portisolation, each port advantageously produces a different gain pattern asshown on FIG. 18B. While this is one specific example, it is understoodthat this structure can be scaled to operate at any desired frequency.It is also understood that methods for tuning, manipulating bandwidth,and creating multiband structures described previously in the context oftwo-port antennas can also apply to this multiport structure.

While the above embodiment is shown as a true cylinder, it is possibleto use other arrangements of three antenna elements and connectingelements that produce the same advantages. This includes, but is notlimited to, arrangements with straight connections such that theconnecting elements form a triangle, or another polygonal geometry. Itis also possible to construct a similar structure by similarlyconnecting three separate dipole elements instead of three monopoleelements with a common counterpoise. Also, while symmetric arrangementof antenna elements advantageously produces equivalent performance fromeach port, e.g., same bandwidth, isolation, impedance matching, it isalso possible to arrange the antenna elements asymmetrically or withunequal spacing depending on the application.

FIG. 19 illustrates use of a multimode antenna structure 1900 in acombiner application in accordance with one or more embodiments of theinvention. As shown in the figure, transmit signals may be applied toboth antenna ports of the antenna structure 1900 simultaneously. In thisconfiguration, the multimode antenna can serve as both antenna and poweramplifier combiner. The high isolation between antenna ports restrictsinteraction between the two amplifiers 1902, 1904, which is known tohave undesirable effects such as signal distortion and loss ofefficiency. Optional impedance matching at 1906 can be provided at theantenna ports.

FIGS. 20A and 20B illustrate a multimode antenna structure 2000 inaccordance with one or more alternate embodiments of the invention. Theantenna structure 2000 can also be used, e.g., in a WiMAX USB orExpressCard/34 device. The antenna structure can be configured foroperation, e.g., in WiMAX bands from 2300 to 6000 MHz.

The antenna structure 2000 includes two antenna elements 2001, 2004,each comprising a broad monopole. A connecting element 2002 electricallyconnects the antenna elements. Slots (or other cut-outs) 2005 are usedto improve the input impedance match above 5000 MHz. The exemplarydesign shown in the figure is optimized to cover frequencies from 2300to 6000 MHz.

The antenna structure 2000 can be manufactured, e.g., by metal stamping.It can be made, e.g., from 0.2 mm thick copper alloy sheet. The antennastructure 2000 includes a pickup feature 2003 on the connecting element2002 generally at the center of mass of the structure, which can be usedin an automated pick-and-place assembly process. The antenna structureis also compatible with surface-mount reflow assembly. Feed points 2006of the antenna provide the points of connection to the radio circuitryon a PCB, and also serve as a support for structural mounting of theantenna to the PCB. Additional contact points 2007 provide structuralsupport.

FIG. 20C illustrates a test assembly 2010 used to measure theperformance of antenna 2000. The figure also shows the coordinatereference for far-field patterns. Antenna 2000 is mounted on a 30×88 mmPCB 2011 representing an ExpressCard/34 device. The grounded portion ofthe PCB 2011 is attached to a larger metal sheet 2012 (having dimensionsof 165×254 mm in this example) to represent a counterpoise size typicalof a notebook computer. Test ports 2014, 2016 on the PCB 2011 areconnected to the antenna through 50-ohm striplines.

FIG. 20D shows the VSWR measured at test ports 2014, 2016. FIG. 20Eshows the coupling (S21 or S12) measured between the test ports. TheVSWR and coupling are advantageously low across the broad range offrequencies, e.g., 2300 to 6000 MHz. FIG. 20F shows the measuredradiation efficiency referenced from the test ports 2014 (Port 1), 2016(Port 2). FIG. 20G shows the calculated correlation between theradiation patterns produced by excitation of test port 2014 (Port 1)versus those produced by excitation of test port 2016 (Port 2). Theradiation efficiency is advantageously high while the correlationbetween patterns is advantageously low at the frequencies of interest.FIG. 20H shows far field gain patterns by excitation of test port 2014(Port 1) or test port 2016 (Port 2) at a frequency of 2500 MHz. FIGS.20I and 20J show the same pattern measurements at frequencies of 3500and 5200 MHz, respectively.

The patterns resulting from test port 2014 (Port 1) are different andcomplementary to those of test port 2016 (Port 2) in the φ=0 or XZ planeand in the θ=90 or XY plane.

FIGS. 21A and 21B illustrate a multimode antenna structure 2100 inaccordance with one or more alternate embodiments of the invention. Theantenna structure 2100 can also be used, e.g., in a WiMAX USB dongle.The antenna structure can be configured for operation, e.g., in WiMAXbands from 2300 to 2400 MHz.

The antenna structure 2100 includes two antenna elements 2102, 2104,each comprising a meandered monopole. The length of the meanderdetermines the center frequency. Other tortuous configurations such as,e.g., helical coils and loops, can also be used to provide a desiredelectrical length. The exemplary design shown in the figure is optimizedfor a center frequency of 2350 MHz. A connecting element 2106 (shown inFIG. 21B) electrically connects the antenna elements 2102, 2104. Atwo-component lumped element match is provided at each antenna feed.

The antenna structure can be fabricated, e.g., from copper as a flexibleprinted circuit (FPC) 2103 mounted on a plastic carrier 2101. Theantenna structure can be created by the metalized portions of the FPC2103. The plastic carrier 2101 provides mounting pins or pips 2107 forattaching the antenna to a PCB assembly (not shown) and pips 2105 forsecuring the FPC 2103 to the carrier 2101. The metalized portion of 2103includes exposed portions or pads 2108 for electrically contacting theantenna to the circuitry on the PCB.

To obtain higher center frequencies, the electrical length of theelements 2102, 2104 can be reduced. FIGS. 22A and 22B illustrate amultimode antenna structure 2200, the design of which is optimized for acenter frequency of 2600 MHz. The electrical length of the elements2202, 2204 is shorter than that of elements 2102, 2104 of FIGS. 21A and21B because metallization at the end of the elements 2202, 2204 has beenremoved, and the width of the of the elements at feed end has beenincreased.

FIG. 23A illustrates a test assembly 2300 using antenna 2100 of FIGS.21A and 21B along with the coordinate reference for far-field patterns.FIG. 23B shows the VSWR measured at test ports 2302 (Port 1), 2304 (Port2). FIG. 23C shows the coupling (S21 or S12) measured between the testports 2302 (Port 1), 2304 (Port 2). The VSWR and coupling areadvantageously low at the frequencies of interest, e.g., 2300 to 2400MHz. FIG. 23D shows the measured radiation efficiency referenced fromthe test ports. FIG. 23E shows the calculated correlation between theradiation patterns produced by excitation of test port 2302 (Port 1)versus those produced by excitation of test port 2304 (Port 2). Theradiation efficiency is advantageously high while the correlationbetween patterns is advantageously low at the frequencies of interest.FIG. 23F shows far field gain patterns by excitation of test port 2302(Port 1) or test port 2304 (Port 2) at a frequency of 2400 MHz. Thepatterns resulting from test port 2302 (Port 1) are different andcomplementary to those of test port 2304 (Port 2) in the φ=0 or XZ planeand in the θ=90 or XY plane.

FIG. 23G shows the VSWR measured at the test ports of assembly 2300 withantenna 2200 in place of antenna 2100. FIG. 23H shows the coupling (S21or S12) measured between the test ports. The VSWR and coupling areadvantageously low at the frequencies of interest, e.g. 2500 to 2700MHz. FIG. 23I shows the measured radiation efficiency referenced fromthe test ports. FIG. 23J shows the calculated correlation between theradiation patterns produced by excitation of test port 2302 (Port 1)versus those produced by excitation of test port 2304 (Port 2). Theradiation efficiency is advantageously high while the correlationbetween patterns is advantageously low at the frequencies of interest.FIG. 23K shows far field gain patterns by excitation of test port 2302(Port 1) or test port 2304 (Port 2) at a frequency of 2600 MHz. Thepatterns resulting from test port 2302 (Port 1) are different andcomplementary to those of test port 2304 (Port 2) in the φ=0 or XZ planeand in the θ=90 or XY plane.

One or more further embodiments of the invention are directed totechniques for beam pattern control for the purpose of null steering orbeam pointing. When such techniques are applied to a conventional arrayantenna (comprising separate antenna elements that are spaced at somefraction of a wavelength), each element of the array antenna is fed witha signal that is a phase shifted version of a reference signal orwaveform. For a uniform linear array with equal excitation, the beampattern produced can be described by the array factor F, which dependson the phase of each individual element and the inter-element elementspacing d.

$F = {A_{0}{\sum\limits_{n = 0}^{N - 1}\; {\exp \left\lbrack {j\; {n\left( {{\beta \; d\; \cos \; \theta} + \alpha} \right)}} \right\rbrack}}}$

where β=2π/λ, N=Total # of elements, α=phase shift between successiveelements, and θ=angle from array axis

By controlling the phase α to a value α_(i), the maximum value of F canbe adjusted to a different direction θ_(i), thereby controlling thedirection in which a maximum signal is broadcast or received.

The inter-element spacing in conventional array antennas is often on theorder of ¼ wavelength, and the antennas can be closely coupled, havingnearly identical polarization. It is advantageous to reduce the couplingbetween elements, as coupling can lead to several problems in the designand performance of array antennas. For example, problems such as patterndistortion and scan blindness (see Stutzman, Antenna Theory and Design,Wiley 1998, pgs 122-128 and 135-136, and 466-472) can arise fromexcessive inter-element coupling, as well as a reduction of the maximumgain attainable for a given number of elements.

Beam pattern control techniques can be advantageously applied to allmultimode antenna structures described herein having antenna elementsconnected by one or more connecting elements, which exhibit highisolation between multiple feedpoints. The phase between ports at thehigh isolation antenna structure can be used for controlling the antennapattern. It has been found that a higher peak gain is achievable ingiven directions when the antenna is used as a simple beam-forming arrayas a result of the reduced coupling between feedpoints. Accordingly,greater gain can be achieved in selected directions from a highisolation antenna structure in accordance with various embodiments thatutilizes phase control of the carrier signals presented to its feedterminals.

In handset applications where the antennas are spaced at much less than¼ wavelength, mutual coupling effects in conventional antennas reducethe radiation efficiency of the array, and therefore reduce the maximumgain achievable.

By controlling the phase of the carrier signal provided to eachfeedpoint of a high isolation antenna in accordance with variousembodiments, the direction of maximum gain produced by the antennapattern can be controlled. A gain advantage of, e.g., 3 dB obtained bybeam steering is advantageous particularly in portable deviceapplications where the beam pattern is fixed and the device orientationis randomly controlled by the user. As shown, e.g., in the schematicblock diagram of FIG. 24, which illustrates a pattern control apparatus2400 in accordance with various embodiments, a relative phase shift α isapplied by a phase shifter 2402 to the RF signals applied to eachantenna feed 2404, 2408. The signals are fed to respective antenna portsof antenna structure 2410.

The phase shifter 2402 can comprise standard phase shift components suchas, e.g., electrically controlled phase shift devices or standard phaseshift networks.

FIGS. 25A-25G provide a comparison of antenna patterns produced by aclosely spaced 2-D conventional array of dipole antennas and a 2-D arrayof high isolation antennas in accordance with various embodiments of theinvention for different phase differences α between two feeds to theantennas. In FIGS. 25A-25G, curves are shown for the antenna patterns atθ=□90 degrees. The solid lines in the figures represents the antennapattern produced by the isolated feed single element antenna inaccordance with various embodiments, while the dashed lines representthe antenna pattern produced by two separate monopole conventionalantennas separated by a distance equal to the width of the singleelement isolated feed structure. Therefore, the conventional antenna andthe high isolation antenna are of generally equivalent size.

In all cases shown in the figures, the peak gain produced by the highisolation antenna in accordance with various embodiments produces agreater gain margin when compared to the two separate conventionaldipoles, while providing azimuthal control of the beam pattern. Thisbehavior makes it possible to use the high isolation antenna in transmitor receive applications where additional gain is needed or desired in aparticular direction. The direction can be controlled by adjusting therelative phase between the drivepoint signals. This may be particularlyadvantageous for portable devices needing to direct energy toward areceive point such as, e.g., a base station. The combined high isolationantenna offers greater advantage when compared to two singleconventional antenna elements when phased in a similar fashion.

As shown in FIG. 25A, the combined dipole in accordance with variousembodiments shows greater gain in a uniform azimuth pattern (θ=90) forα=0 (zero degrees phase difference).

As shown in FIG. 25B, the combined dipole in accordance with variousembodiments shows greater peak gain (at φ=0) with a non-symmetricazimuthal pattern (θ=90 plot□ for α=30 (30 degrees phase differencebetween feedpoints).

As shown in FIG. 25C, the combined dipole in accordance with variousembodiments shows greater peak gain (at φ=0) with a shifted azimuthalpattern (θ=90 plot□ for α=60 (60 degrees phase difference betweenfeedpoints).

As shown in FIG. 25D, the combined dipole in accordance with variousembodiments shows even greater peak gain (at φ=0) with a shiftedazimuthal pattern (θ=90 plot□ for α=90 (90 degrees phase differencebetween feedpoints).

As shown in FIG. 25E, the combined dipole in accordance with variousembodiments shows greater peak gain (at φ=0) with a shifted azimuthalpattern (θ=90 plot greater backlobe (at φ=180) for α=120 (120 degreesphase difference between feedpoints).

As shown in FIG. 25F, the combined dipole in accordance with variousembodiments shows greater peak gain (at φ=0) with a shifted azimuthalpattern (θ=90 plot), even greater backlobe (at φ=180) for α=150 (150degrees phase difference between feedpoints).

As shown in FIG. 25G, the combined dipole in accordance with variousembodiments shows greater peak gain (at φ=0 & 180) with a double lobedazimuthal pattern (θ=90 plot) for α=180 (180 degrees phase differencebetween feedpoints).

FIG. 26 illustrates the ideal gain advantage if the combined highisolation antenna in accordance with one or more embodiments over twoseparate dipoles as a function of the phase angle difference between thefeedpoints for a two feedpoint antenna array.

The increased gain obtained by pattern control using an antennastructure with two parallel dipoles connected by meandered connectingelements in accordance with one or more embodiments of the invention maybe utilized to improve the range or reliability of a wireless link.Alternately, the increased gain may allow for a portable or other deviceto obtain equivalent wireless link performance with reduced transmitpower. For example, an average transmit gain improvement of 3 dBobtained from pattern control would allow for the transmit power to bereduced by 3 dB while maintaining the same link performance. Reductionof transmit power is advantageous in several ways. First, portablewireless devices are typically required to meet a specific absorptionrate (SAR) regulatory limit, which can be difficult to meet without someperformance compromise. A reduction in transmit power can provide acorresponding reduction in the peak SAR value without performancecompromise. In addition, lower transmit power reduces the burden on theoutput PA, allowing design for lower power and higher linearity.Furthermore, reduced transmit power is beneficial for longer batterylife and lower heat dissipation requirements for portable or otherdevices.

While the use of phase control produces a desired increase in far-fieldgain, changes in phase excitation may also alter the near-fields andaffect SAR values. To realize a net SAR value reduction, the antennafar-field gain increase should be greater than any increase in peak SARvalue. Through experimentation, Applicants have found that in fact thechange in SAR value is relatively small over phase in comparison to thefar-field gain.

An exemplary USB dongle with two port antenna structure for patterncontrol application in the 1900 MHz band is shown on FIG. 30. As shownin FIG. 31, the SAR value, as determined by simulation for theconfiguration of FIG. 30, is relatively independent of the relativephase between the drivepoint signals used for pattern control, so thatthe benefit of the reduction in measured peak SAR value is achievablefor all relative phase values, while providing full azimuthal control ofthe beam pattern.

The techniques described herein for reducing near-field radiation levelsand SAR values are preferably used with the high isolation multimodeantenna structures described above having connecting elementselectrically connecting the antenna elements. However, the techniquescan also be more generally used with antenna arrays comprising aplurality of radiating elements that are phase steerable to provideantenna pattern control and to increase gain in a selected direction.

Further embodiments of the invention are directed to multimode antennastructures that provide increased high isolation between multi-bandantenna ports operating in close proximity to each other at a givenfrequency range. In these embodiments, a band-rejection slot isincorporated in one of the antenna elements of the antenna structure toprovide reduced coupling at the frequency to which the slot is tuned.

FIG. 27A schematically illustrates a simple dual-band branch linemonopole antenna 2700. The antenna 2700 includes a band-rejection slot2702, which defines two branch resonators 2704, 2706. The antenna isdriven by signal generator 2708. Depending on the frequency at which theantenna 2700 is driven, various current distributions are realized onthe two branch resonators 2704, 2706.

The physical dimensions of the slot 2702 are defined by the width Ws andthe length Ls as shown in FIG. 27A. When the excitation frequencysatisfies the condition of Ls=lo/4, the slot feature becomes resonant.At this point the current distribution is concentrated around theshorted section of the slot, as shown in FIG. 27B.

The currents flowing through the branch resonators 2704, 2706 areapproximately equal and oppositely directed along the sides of the slot2702. This causes the antenna structure 2700 to behave in a similarmanner to a spurline band stop filter 2720 (shown schematically in FIG.27C), which transforms the antenna input impedance down significantlylower than the nominal source impedance. This large impedance mismatchresults in a very high VSWR, shown in FIG. 27D and 27E, and as a resultleads to the desired frequency rejection.

This band-rejection slot technique can be applied to an antenna systemwith two (or more) antennas elements operating in close proximity toeach other where one antenna element needs to pass signals of a desiredfrequency and the other does not. In one or more embodiments, one of thetwo antenna elements includes a band-rejection slot, and the other doesnot. FIG. 28 schematically illustrates an antenna structure 2800, whichincludes a first antenna element 2802, a second antenna element 2804,and a connecting element 2806. The antenna structure 2800 includes ports2808 and 2810 at antenna elements 2802 and 2804, respectively. In thisexample, a signal generator drives the antenna structure 2802 at port2808, while a meter is coupled to the port 2810 to measure current atport 2810. It should be understood, however, that either or both portscan be driven by signal generators. The antenna element 2802 includes aband-rejection slot 2812, which defines two branch resonators 2814,2816. In this embodiment, the branch resonators comprise the maintransmit section of the antenna structure, while the antenna element2804 comprises a diversity receive portion of the antenna structure.

Due to the large mismatch at the port of the antenna element 2802 withthe band-reject slot 2812, the mutual coupling between it and thediversity receive antenna element 2804, which is actually matched at theslot resonant frequency will be quite small and will result inrelatively high isolation.

FIG. 29A is a perspective view of a multimode antenna structure 2900comprising a multi-band diversity receive antenna system that utilizesthe band-rejection slot technique in the GPS band in accordance with oneor more further embodiments of the invention. (The GPS band is 1575.42MHz with 20 MHz bandwidth.) The antenna structure 2900 is formed on aflex film dielectric substrate 2902, which is formed as a layer on adielectric carrier 2904. The antenna structure 2900 includes a GPS bandrejection slot 2906 on the primary transmit antenna element 2908 of theantenna structure 2900. The antenna structure 2900 also includes adiversity receive antenna element 2910, and a connecting element 2912connecting the diversity receive antenna element 2910 and the primarytransmit antenna element 2908. A GPS receiver (not shown) is connectedto the diversity receive antenna element 2910. In order to generallyminimize the antenna coupling from the primary transmit antenna element2908 and to generally maximize the diversity antenna radiationefficiency at these frequencies, the primary antenna element 2908includes the band-rejection slot 2906 and is tuned to an electricalquarter wave length near the center of the GPS band. The diversityreceive antenna element 2910 does not contain such a band rejectionslot, but comprises a GPS antenna element that is properly matched tothe main antenna source impedance so that there will be generallymaximum power transfer between it and the GPS receiver. Although bothantenna elements 2908, 2910 co-exist in close proximity, the high VSWRdue to the slot 2906 at the primary transmit antenna element 2908reduces the coupling to the primary antenna element source resistance atthe frequency to which the slot 2906 is tuned, and therefore providesisolation at the GPS frequency between both antenna elements 2908, 2910.The resultant mismatch between the two antenna elements 2908, 2910within the GPS band is large enough to decouple the antenna elements inorder to meet the isolation requirements for the system design as shownin FIGS. 29B and 29C.

In the antenna structures described herein in accordance with variousembodiments of the invention, the antenna elements and the connectingelements preferably form a single integrated radiating structure suchthat a signal fed to either port excites the entire antenna structure toradiate as a whole, rather than separate radiating structures. As such,the techniques described herein provide isolation of the antenna portswithout the use of decoupling networks at the antenna feed points

It is to be understood that although the invention has been describedabove in terms of particular embodiments, the foregoing embodiments areprovided as illustrative only, and do not limit or define the scope ofthe invention.

Various other embodiments, including but not limited to the following,are also within the scope of the claims. For example, the elements orcomponents of the various multimode antenna structures described hereinmay be further divided into additional components or joined together toform fewer components for performing the same functions.

Having described preferred embodiments of the present invention, itshould be apparent that modifications can be made without departing fromthe spirit and scope of the invention.

What is claimed is:
 1. A method, comprising: introducing a phase offsetbetween signals applied to antenna ports of an antenna structure suchthat a first signal applied to one of the antenna ports has a differentphase than a second signal applied to another one of the antenna portsto provide antenna pattern control and to increase gain in a directiontoward a far-field point, wherein the antenna structure comprises aplurality of antenna elements, each operatively coupled to a differentone of the antenna ports, wherein electrical currents flowing in theantenna structure are such that the antenna ports are isolated at aparticular signal frequency range; and operating at a reduced power thatis lower than a power used in a non-pattern control operation of theantenna structure such that a first communications device utilizing theantenna structure satisfies an equivalent wireless link performance witha second communications device at the far-field point using the reducedpower compared to the non-pattern control operation, thereby reducingnear field radiation.
 2. The method of claim 1, wherein the introducingof the phase offset between the signals comprises adjusting a relativephase between the signals using an electrically controlled phase shiftdevice.
 3. The method of claim 1, wherein the introducing of the phaseoffset between the signals comprises adjusting a relative phase betweenthe signals using a phase shift network.
 4. The method of claim 1,wherein the introducing of the phase offset between the signalscomprises adjusting a relative phase between the signals by controllinga phase of a carrier signal provided at each of the antenna ports. 5.The method of claim 1, wherein the first communications device is acellular handset, personal digital assistance, wireless access point, ora data card for a computer.
 6. The method of claim 1, wherein theplurality of antenna elements comprise helical coils, wideband planershapes, chip antennas, meandered shapes, loops, or inductively shuntedforms.
 7. The method of claim 1, wherein the antenna structure comprisesa planar structure fabricated on a printed circuit board substrate. 8.The method of claim 1, wherein the antenna structure comprises stampedmetal part including a pickup feature for use in a pick and placeassembly process.
 9. The method of claim 1, wherein the antennastructure comprises a flexible printed circuit mounted on a plasticcarrier or on a plastic housing of a device.
 10. The method of claim 1,wherein the second communications device comprises a base station, amobile terminal, or a router.
 11. The method of claim 1, wherein theintroducing of the phase offset between the signals comprises adjustinga relative phase between signals fed to antenna ports operativelycoupled to neighboring antenna elements to maintain a communicationslink with equipment at the far-field point.
 12. A method, comprising:introducing a phase offset between signals applied to antenna ports ofan antenna structure comprising a plurality of antenna elements, suchthat a first signal applied to one of the antenna ports operativelycoupled to one of the plurality of antenna elements has a differentphase than a second signal applied to another one of antenna portsoperatively coupled to another one of the plurality of antenna elementsto provide antenna pattern control and to increase gain in a selecteddirection toward a far-field point; and operating at a first power levelthat is lower than a second power used in a non-pattern controloperation of the antenna structure to meet wireless link performancecriteria with equipment at the far-field point using reduced powercompared to the non-pattern control operation, thereby reducing nearfield radiation.
 13. The method of claim 12, wherein the introducing ofthe phase offset between the signals comprises adjusting a phase betweenthe signals using an electrically controlled phase shift device.
 14. Themethod of claim 12, wherein the introducing of the phase offset betweenthe signals comprises adjusting the phase between the signals using aphase shift network.
 15. The method of claim 12, wherein the introducingof the phase offset between the signals comprises adjusting the phasebetween the signals by controlling the phase of a carrier signalprovided at each of the antenna ports.
 16. The method of claim 12,wherein the antenna structure is utilized in a communication device, andwherein the communications device is a cellular handset, personaldigital assistance, wireless access point, or a data card for computer.17. The method of claim 12, wherein the equipment at the far-field pointcomprises a base station, a mobile terminal, or a router.
 18. A method,comprising: introducing a phase offset between signals applied toantenna ports of an antenna comprising a plurality of antenna elements,such that a first signal applied to one of the antenna ports operativelycoupled to one of the plurality of antenna elements has a differentphase than a second signal applied another one of antenna portsoperatively coupled to another one of the plurality of antenna elementsto obtain an antenna pattern control; and using a reduced power that islower than the power used in a non-pattern control operation of theantenna such that a wireless link performance criteria is met withequipment at a far-field point using the reduced power compared to thenon-pattern control operation, thereby reducing near field radiation.19. The method of claim 18, wherein the equipment at the far-field pointcomprises a base station, a mobile terminal, or a router.
 20. The methodof claim 18, wherein the introducing of the phase offset between thesignals comprises adjusting the phase between the signals using a devicethat controls phase.